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| NCP1351 Product Preview Variable Off Time PWM Controller The NCP1351 is a current-mode controller targeting low power off-line flyback Switched Mode Power Supplies (SMPS) where cost is of utmost importance. Based on a fixed peak current technique (quasi-fixed TON), the controller decreases its switching frequency as the load becomes lighter. As a result, a power supply using the NCP1351 naturally offers excellent no-load power consumption, while optimizing the efficiency in other loading conditions. When the frequency decreases, the peak current is gradually reduced down to approximately 30% of the maximum peak current to prevent transformer mechanical resonance. The risk of acoustic noise is thus greatly diminished while keeping good standby power performance. An externally adjustable timer permanently monitors the feedback activity and protects the supply in presence of a short-circuit or an overload. Once the timer elapses, NCP1351 stops switching and stays latched for version A, and tries to restart for Version B. The internal structure features an optimized arrangement which allows one of the lowest available startup current, a fundamental parameter when designing low standby power supplies. The negative current sensing technique minimizes the impact of the switching noise on the controller operation and offers the user to select the maximum peak voltage across his current sense resistor. Its power dissipation can thus be application optimized. Finally, the bulk input ripple ensures a natural frequency smearing which smooths the EMI signature. Features http://onsemi.com MARKING DIAGRAM 8 8 1 SOIC-8 D SUFFIX CASE 751 1 x A Y WW G 1351x AYWW G = A, B, C, or D Options = Assembly Location = Year = Work Week = Pb-Free Device PIN CONNECTIONS FB 1 Ct 2 CS 3 GND 4 (Top View) 8 TIMER 7 LATCH 6 VCC 5 DRV ORDERING INFORMATION Device NCP1351ADR2G Package Shipping * * * * * * * * * * * * Quasi-fixed TON, Variable TOFF Current Mode Control Extremely Low Current Consumption at Startup Peak Current Compression Reduces Transformer Noise Primary or Secondary Side Regulation Dedicated Latch Input for OTP, OVP Programmable Current Sense Resistor Peak Voltage Natural Frequency Dithering for Improved EMI Signature Easy External Over Power Protection (OPP) Undervoltage Lockout Very Low Standby Power via Off-time Expansion Internal Temperature Shutdown SOIC-8 Package SOIC-8 2500 / Tape & Reel (Pb-Free) SOIC-8 2500 / Tape & Reel (Pb-Free) NCP1351BDR2G For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging Specifications Brochure, BRD8011/D. Typical Applications * Auxiliary Power Supply * Printer, Game Stations, Low-Cost Adapters * Off-line Battery Charger This document contains information on a product under development. ON Semiconductor reserves the right to change or discontinue this product without notice. (c) Semiconductor Components Industries, LLC, 2006 1 November, 2006 - Rev. P2 Publication Order Number: NCP1351/D NCP1351 + VOUT NCP1351 + 85-265VAC *OPP 1 2 3 4 8 7 6 5 LATCH + GND *Optional Figure 1. Typical Application Circuit PIN FUNCTION DESCRIPTION Pin N 1 2 3 4 5 6 7 8 Pin Name FB Ct CS GND DRV VCC Latch Timer Function Feedback Input Oscillator Frequency Current Sense Input - Driver Output Supply Input Latchoff Input Fault Timer Capacitor Pin Description Injecting Current in this Pin Reduces Frequency A capacitor sets the maximum switching frequency at no feedback current Senses the Primary Current - Driving Pulses to the Power MOSFET Supplies the controller up to 28 V A positive voltage above VLATCH fully latches off the controller Sets the time duration before fault validation http://onsemi.com 2 NCP1351 INTERNAL CIRCUIT ARCHITECTURE VDD 20 ms Filter VTIMER FB TIMER UVLO Reset Fault = Low + VDD + - VFault IP Flag 20 ms Filter ICt S Ct Q Q UVLO Reset 45k R VDD VCC Mngt VZENER VCCSTOP 1 = OK 0 = not OK Clamp 4V Reset VDD VOFFset + 1 ms Pulse ICS-dif* S ICS-dif* Q Q CS ICS-min* R GND Vth + - + *(ICS-diff = ICS-max -ICS-min) Figure 2. A Version (Latched Short-Circuit Protection) http://onsemi.com 3 - + + VDD ITIMER - + + VLATCH LATCH VCC + - DRV NCP1351 VDD S Q VITIMER FB Q UVLO Reset R Fault = Low + VDD + - VFault IP Flag 20 ms Filter ICt S Ct Q Q UVLO Reset 45k R VDD VCC Mngt VZENER VCCSTOP 1 = OK 0 = not OK Clamp 4V Reset VDD VOFFset + 1 ms Pulse ICS-dif* S ICS-dif* Q Q CS ICS-min* R GND Vth + - + *(ICS-diff = ICS-max -ICS-min) Figure 3. B Version (Auto-recovery Short-Circuit Protection) http://onsemi.com 4 - + + VDD ITIMER TIMER - + + VLATCH LATCH VCC + - DRV NCP1351 MAXIMUM RATINGS Symbol VSUPPLY ISUPPLY VDRV IDRV VMAX IMAX IFBmax RGmin RqJA TJMAX Maximum Supply on VCC Pin 6 Maximum Current in VCC Pin 6 Maximum Voltage on DRV Pin 5 Maximum Current in DRV Pin 5 Supply Voltage on all pins, except Pin 6 (VCC), Pin 5 (DRV) Maximum Current in all Pins Except Pin 6 (VCC) and Pin 5 (DRV) Maximum Injected Current in Pin 1 (FB) Minimum Resistive Load on DRV Pin Thermal Resistance Junction-to-Air Maximum Junction Temperature Storage Temperature Range ESD Capability, Human Body Model V per Mil-STD-883, Method 3015 ESD Capability, Machine Model Rating Value -0.3 to 28 20 -0.3 to 20 $400 -0.3 to 10 $10 0.5 33 200 150 -60 to +150 2 200 Unit V mA V mA V mA mA kW C/W C C kV V Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect device reliability. NOTE: This device contains latchup protection and exceeds 100 mA per JEDEC Standard JESD78. http://onsemi.com 5 NCP1351 Electrical Characteristics (For typical values TJ = 25C, for Min/Max Values TJ = -25C to +125C, Max TJ = 150C, VCC = 12 V unless otherwise noted) Symbol SUPPLY SECTION AND VCC MANAGEMENT VCCON VCCSTOP VCCHYST VZENER ICC1 ICC2 ICC3 ICCLATCH VCC Increasing Level at Which Driving Pulses are Authorized VCC Decreasing Level at Which Driving Pulses are Stopped Hysteresis VccON - VccSTOP Clamped VCC When Latched off / Burst Mode Activation Startup Current Internal IC Consumption with IFB = 50 mA, FSW = 65 kHz and CL = 0 Internal IC Consumption with IFB = 50 mA, FSW = 65 kHz and CL = 1 nF Current Flowing into VCC pin that Keeps the Controller Latched 6 6 6 6 6 6 6 6 15 8.3 6 - - - - 20 18 8.9 - 6 - 1.0 1.6 - 22 9.5 - - 10 1.8 2.5 - V V V V mA mA mA mA Rating Pin Min Typ Max Unit CURRENT SENSE ICSmin ICSmin ICSmax ICSmax VTH tdelay Minimum Source Current (IFB = 90 mA) Minimum Source Current (IFB = 90 mA) Maximum Source Current (IFB = 50 mA) Maximum Source Current (IFB = 50 mA) Current Sense Comparator Threshold Voltage Propagation Time Delay (CS Falling Edge to Gate Output) TJ = 0C to +125C TJ = -25C to +125C TJ = 0C to +125C TJ = -25C to +125C 3 3 3 3 3 3 61 58 251 242 10 - 70 70 270 270 20 160 75 75 289 289 35 300 mA mA mA mA mV ns TIMING CAPACITOR VOFFSET VCTMAX ICT VCTMIN TDISCH VFAULT Minimum Voltage on CT Capacitor, IFB = 30 mA Voltage on CT Capacitor at IFB = 150 mA Source Current (Ct Pin Grounded) Minimum Voltage on CT, Discharge Switch Activated CT Capacitor Discharge Time (Activated at DRV Turn-on) CT Capacitor Level at Which Fault Timer Starts A, B Versions 2 2 2 2 2 2 0.4 475 5 10 - 510 - 11 - 1 0.5 0.6 565 - 12 20 mV V mA mV ms V FEEDBACK SECTION VFB IFAULT IFBcomp IFBred FB Pin Voltage for an Injected Current of 200 mA FB Current Under Which a Fault is Detected FB Current at Which CS Compression Starts FB Current at Which CS Compression is Finished A, B versions 1 1 1 1 - - - - 0.7 40 60 80 - - - - V mA mA mA Drive Output Tr Tf ROH ROL VDRVlow VDRVhigh Protection ITIMER VTIMER TTIMER VLATCH Timing Capacitor Charging Current Fault Voltage on Pin 8 Fault Timer Duration, CTIMER = 100 nF Latching Voltage 8 8 - 7 10 4.5 - 4.5 11.5 5 42 5 13 5.5 - 5.5 mA V ms V Output Voltage Rise-time @ CL = 1 nF, 10 - 90% of Output Signal Output Voltage Fall-time @ CL = 1 nF, 10 - 90% of Output Signal Source Resistance Sink Resistance DRV Pin Level at VCC Close to VCCSTOP with a 33 kW Resistor to GND DRV Pin Level at VCC = 28 V 5 5 5 5 5 5 - - - - 8.0 16 90 100 80 30 - 17 - - - - - 20 ns ns W W V V http://onsemi.com 6 NCP1351 The NCP1351 implements a fixed peak current mode technique whose regulation scheme implements a variable switching frequency. As shown on the typical application diagram, the controller is designed to operate with a minimum number of external components. It incorporates the following features: * Frequency Foldback: Since the switching period increases when power demand decreases, the switching frequency naturally diminishes in light load conditions. This helps to minimize switching losses and offers good standby power performance. * Very Low Startup Current: The patented internal supply block is specially designed to offer a very low current consumption during startup. It allows the use of a very high value external startup resistor, greatly reducing dissipation, improving efficiency and minimizing standby power consumption. * Natural Frequency Dithering: The quasi-fixed tON mode of operation improves the EMI signature since the switching frequency varies with the natural bulk ripple voltage. * Peak Current Compression: As the load becomes lighter, the frequency decreases and can enter the audible range. To avoid exciting transformer mechanical resonances, hence generating acoustic noise, the NCP1351 includes a patented technique, which reduces the peak current as power goes down. As such, inexpensive transformer can be used without having noise problems. * Negative Primary Current Sensing: By sensing the total current, this technique does not modify the MOSFET driving voltage (VGS) while switching. Furthermore, the programming resistor, together with the pin capacitance, forms a residual noise filter which blanks spurious spikes. Programmable Primary Current Sense: It offers a second peak current adjustment variable, which improves the design flexibility. Extended VCC Range: By accepting VCC levels up to 28 V, the device offers added flexibility in presence of loosely coupled transformers. The gate drive is safely clamped below 20 V to avoid stressing the driven MOSFET. Easy OPP: Connecting a resistor from the CS pin to the auxiliary winding allows easy bulk voltage compensation. Secondary or Primary Regulation: The feedback loop arrangement allows simple secondary or primary side regulation without significant additional external components. Latch Input: If voltage on Pin 7 is externally brought above 5 V, the controller permanently latches off and stays latched until the user cycles VCC down, below 4 V typically. Fault Timer: In presence of badly coupled transformer, it can be quite difficult to detect an overload or a short-circuit on the primary side. When the feedback current disappears, a current source charges a capacitor connected to Pin 8. When the voltage on this pin reaches a certain level, all pulses are shut off and the VCC voltage is pulled down below the VCC(min) level. This protection is latched on the A version (the controller must be shut down and restart to resume normal operation), and auto-recovery on Version B (if the fault goes away, the controller automatically resumes operation). * * * * * * http://onsemi.com 7 NCP1351 APPLICATION INFORMATION The Negative Sensing Technique Standard current-mode controllers use the positive sensing technique as portrayed by Figure 4. In this technique, the controller detects a positive voltage drop across the sense resistor, representative of the flowing current. Unfortunately, this solution suffers from the following drawbacks: 1. Difficulties to precisely adjust the peak current. If 1 V is the maximum sense level, you must combine low valued resistors to reach the exact limit you need. 2. The voltage developed across the sense resistor subtracts from the gate voltage. If your VCC(min) is 7 V, then the actual gate voltage at the end of the on time, assuming a full load condition, is 7 V - 1 V = 6 V. 3. The current in the sense resistor also includes the Ciss current at turn-on. This narrow spike often disturbs the controller and requires adequate treatment through a LEB circuitry for instance. Figure 5 represents the negative current sense technique. In this simplified example, the source directly connects to the controller ground. Hence, if VCC is 8 V, the effective gate-source voltage is very close to 8 V: no sense resistor drop. How does the controller detect a negative excursion? In lack of primary current, the voltage on the CS pin reaches Roffset x ICS. Let us assume that these elements lead to have 1 V on this pin. Now, when the power MOSFET activates, the current flows via the sense resistor and develop a negative voltage by respect to the controller ground. The voltage seen on the CS is nothing else than a positive voltage (Roffset x ICS) plus the voltage across the sense resistor which is negative. Thus, the CS pin voltage goes low as the primary ILp current increases. When the result reaches the threshold voltage (around 20 mV), the comparator toggles and resets the main latch. Figure 3 details how the voltage moves on the CS pin on a 1351 demoboard, whereas Figure 7 zooms on the sense resistor voltage captured by respect to the controller ground. The choice of these two elements is simple. Suppose you want to develop 1 V across the sense resistor. You would select the offset resistor via the following formula: Roffset + 1 ICS + 1 + 3.7 kW 270 m (eq. 1) If you need a peak current of 2 A, then, simply apply the ohm law to obtain the sense resistor value: Rsense + 1 Ipeak_max + 1 + 0.5 W 2 (eq. 2) Due to the circuit flexibility, suppose you only have access to a 0.33 W resistor. In that case, the peak current will exceed the 2 A limit. Why not changing the offset resistor value then? To obtain 2 A from the 0.33 W resistor, you should develop: The offset resistor is thus derived by: Vsense + RsenseIpeak_max + 0.33 0.66 ICS 2 + 660 mV (eq. 3) Roffset + + 0.66 + 2.44 kW 270 m (eq. 4) If reducing the sense resistor is of good practice to improve the efficiency, we recommend to adopt sense values between 0.5 V and 1 V. Reducing the voltage below these levels will degrade the noise immunity. LP LP DRV VDD + + DRV CBulk CS ILp Reset + - Vgs ILp CBulk ILp ICS CS - + Reset GND ILp Peak Setpoint Rsense GND ILp Vsense Voffset Roffset + Vth ILp Vsense Figure 4. Positive Current-Sense Technique Figure 5. A Simplified Circuit of the Negative Sense Implementation http://onsemi.com 8 NCP1351 Current Sense Resistor Current Sense Pin Figure 6. The Voltage on the Current Sense Pin Figure 7. The Voltage Across the Sense Resistor Below are a few recommendations concerning the wiring and the PCB layout: * A small 22 pF capacitor can be placed between the CS pin and the controller ground. Place it as close as possible to the controller. * Do not place the offset resistor in the vicinity of the sense element, but put it close to the controller as well. * Regulation by frequency * The power a flyback converter can deliver relates to the energy stored in the primary inductance Lp and obeys the following formulae: Pout_DCM + 1 LP Ipeak2 FSW h 2 (eq. 5) (eq. 6) Pout_CCM + 12 LP(Ipeak2 * Ivalley2)FSW h Where: (eta) is the converter efficiency Ipeak is the peak inductor current reached at the on time termination Ivalley represents the current at the end of the off time. It equals zero in DCM. FSW is the operating frequency. VCt Thus, to control the delivered power, we can either play on the peak current setpoint (classical peak current mode control) or adjust the switching frequency by keeping the peak current constant. We have chosen the second scheme in this NCP1351 for simplicity and ease of implementation. Thus, once the peak current has been selected, the feedback loop automatically reacts to satisfy Equations 5 and 6. The external capacitor that you connect between pin 2 and ground (again, place it close to the controller pins) sets the maximum frequency you authorize the converter to operate up to. Normalized values for this timing capacitor are 270 pF (65 kHz) and 180 pF (100 kHz). Of course, different combinations can be tried to design at higher or lower frequencies. Please note that changing the capacitor value does not affect the operating frequency at nominal line and load conditions. Again, the operating frequency is selected by the feedback loop to cope with Equations 5 and 6 definitions. The feedback current controls the frequency by changing the timing capacitor end of charge voltage, as illustrated by Figure 8. Controlled by the FB Current Minimum Frequency Pout Decreases Pout Increases ICt = 10 mA Maximum Frequency Figure 8. The Current Injected into the Feedback Loop Adjusts the Switching Frequency http://onsemi.com 9 NCP1351 Ct Voltage Ct Voltage Figure 9. In Light Load Conditions, the Oscillator Further Delays the Restart Time Figure 10. Ct Voltage Swing at a Moderate Loading In light load conditions, the frequency can go down to a few hundred Hz without any problem. The internal circuitry naturally blocks the oscillator and softly shifts the restart time as shown on Figure 9 scope shot. Delays The Restart Time D1 1N4148 DRV Q1 2N2907 GND In lack of feedback current, for instance during a startup sequence or a short circuit, the oscillator frequency is pushed to the limit set by the timing capacitor. In this case, the lower threshold imposed to the timing capacitor is blocked to 500 mV (parameter Vfault). This is the maximum power the converter can deliver. To the opposite, as you inject current via the optocoupler in the feedback pin, the off time expands and the power delivery reduces. The maximum threshold level in standby conditions is set to 6 V. Over Power Protection Figure 11. A Low-Cost PNP Improves the Drive Capability at Turn-off As any universal-mains operated converters, the output power slightly increases at high line compared to what the power supply can deliver at low line. This discrepancy relates to the propagation delay from the point where the peak is detected to the MOSFET gate effective pulldown. It naturally includes the controller reaction time, but also the driver capability to pull the gate down. If the MOSFET Qg is too large, then this parameter will greatly affect your overpower parameter. Sometimes, the small PNP can help and we recommend it if you use a large Qg MOSFET: Over power protection can be done without power dissipation penalty by arranging components around the auxiliary as suggested by Figure 11. On this schematic, the diode anode swings negative during the on time. This negative level directly depends on the input voltage and offsets the current sense pin via the ROPP resistor. A small integration is necessary to reduce the OPP action in light load conditions. However, depending on the compensation level, the standby power can be affected. Again, the resistor ROPP should be placed as close as possible to the CS pin. The 22 pF can help to circumvent any picked-up noise and D2 prevents the positive loading of the 270 pF capacitor during the flyback swing. We have put a typical 100 kW OPP resistor but a tweak is required depending on your application. http://onsemi.com 10 NCP1351 LP DRV + DRV Daux + CS C4 22p VCC CBulk ILp CVCC Laux R1 150k C3 270p Roffset Rsense D2 1N4148 ROPP 100k Figure 12. The OPP is Relatively Easy to Implement and It Does not Waste Power Suppose you would need to reduce the peak current by 15% in high-line conditions. The turn-ratio between the auxiliary winding and the primary winding is Naux. Assume its value is 0.15. Thus, the voltage on Daux cathode swings negative during the on time to a level of: Vaux_peak + -Vin_max Naux + -375 0.15 + -56 V (eq. 7) 15%. Compared to the internal 270 mA source, we need to derive: Ioffset + -0.15 270 m + -40.5 mA (eq. 10) Thus, from the -4 V excursion, the ROPP resistor is derived by: ROPP + 4 + 98 kW 40.5 m (eq. 11) If we selected a 3.7 kW resistor for Roffset, then the maximum sense voltage being developed is: Vsense + 3.7 k 270 m + 1 V (eq. 8) After experimental measurements, the resistor was normalized down to 100 kW. Feedback The small RC network made of R1 and C3, purposely limits the voltage excursion on D2 anode. Assume the primary inductance value gives an on time of 3 s at high-line. The voltage across C3 thus swings down to: VC3 + tonVaux_peak R1C3 +- 3m 150 k 56 270 p + -4.2 V (eq. 9) Unlike other controllers, the feedback in the NCP1351 works in current rather than voltage. Figure 13 details the internal circuitry of this particular section. The optocoupler injects a current into the FB pin in relationship with the input/output conditions. Typically, we measured around -4 V on our 50 W prototype. By calculation, we want to decrease the peak current by http://onsemi.com 11 NCP1351 VCC ICt 10m Ct Ct 270p VCC Reset FB IFB IFB Voffset 500mV + - Clock + C1 100n R1 2.5k DFB IFB RFB 45k VCC C3 22pF ICSmin CS Idiff = ICSmax - ICSmin Idiff Idiff Roffset 3.9k f(IFB) to Rsense Figure 13. The Feedback Section Inside the NCP1351 The FB pin can actually be seen as a diode, forward biased by the optocoupler current. The feedback current, IFB on Figure 13, enter an internal 45 kW resistor which develops a voltage. This voltage becomes the variable threshold point for the capacitor charge, as indicated by Figure 8. Thus, in lack of feedback current (start-up or short-circuit), there is no voltage across the 45 kW and the series offset of 500 mV clamps the capacitor swing. If a 270 pF capacitor is used, the maximum switching frequency is 65 kHz. Folding the frequency back at a rather high peak current can obviously generate audible noise. For this reason, the NCP1351 uses a patented current compression technique which reduces the peak current in lighter load conditions. By design, the peak current changes from 100% of its full load value, to 30% of this value in light load conditions. This is the block placed on the lower left corner of Figure 13. In full load conditions, the feedback current is weak and all the current flowing through the external offset resistor is: ICS + ICS_min ) Idif + ICS_max * ICS_min + ICS_max (eq. 12) As the load goes lighter, the feedback current increases and starts to steal current away from the generators. Equation 12 can thus be updated by: ICS + ICS_max * kIFB (eq. 13) Equation 13 testifies for the current reduction on the offset generator, k represents an internal coefficient. When the feedback current equals Idif, the offset becomes: ICS + ICS_min (eq. 14) http://onsemi.com 12 NCP1351 At this point, the current is fully compressed and remains frozen. To further decrease the transmitted power, the frequency does not have other choice than going down. CS Current 250 mA R1 2.5k FAULT (A, B versions) C1 100nF VCC FB 70 mA Figure 15. The Recommended Feedback Arrangement Around the FB Pin 40 mA 60 mA 80 mA Fault detection FB Current Figure 14. The NCP1351 Peak Current Compression Scheme Looking to the data-sheet specifications, the maximum peak current is set to 270 mA whereas the compressed current goes down to 70 mA. The NCP1351 can thus be considered as a multi operating mode circuit: * Real fixed peak current / variable frequency mode for FB current below 60 mA. * Then maximum peak current decreases to ICS,min over a narrow linear range of IFB (to avoid instability created by a discrete jump from ICS,max to ICS,min), between 60 mA and 80 mA. * Then if IFB keeps on increasing, in a real fixed peak current/variable frequency mode with reduced peak current For biasing purposes and noise immunity improvements, we recommend to wire a pulldown resistor and a capacitor in parallel from the FB pin to the controller ground (Figure 15). Please keep these elements as close as possible to the circuit. The pulldown resistor increases the optocoupler current but also plays a role in standby. We found that a 2.5 kW resistor was giving a good tradeoff between optocoupler operating current (internal pole position) and standby power. The fault detection circuitry permanently observes the FB current, as shown on Figure 17. When the feedback current decreases below 40 mA, an external capacitor is charged by a 11.7 mA source. As the voltage rises, a comparator detects when it reaches 5 V typical. Upon detection, there can be two different scenarios: 1. A version: the circuit immediately latches-off and remains latched until the voltage on the current into the VCC pin drops below a few A. The latch is made via an internal SCR circuit who holds Vcc to around 6 V when fired. As long as the current flowing through this latch is above a few mA, the circuit remains locked-out. When the user unplugs the converter, the VCC current falls down and resets the latch. 2. B version: the circuit stops its output pulses and the auxiliary VCC decreases via the controller own consumption (600 mA). When it touches the VCC(min) point, the circuit re-starts and attempts to crank the power supply. If it fails again, an hiccup mode takes place (Figure 13). VCC Vdrv Figure 16. Hiccup Occurs with the B Version Only, the A Version Being Latched The duty-burst in fault is around 7% in this particular case. http://onsemi.com 13 NCP1351 VCC Itimer 10m Timer Ctimer 100nF VCC Pon Reset IFB DFB IFB IFB + + VCC + - S Q Q R - + VCC == VCC(min) ? Reset DRV Pulses to DRV Stage 20ms Filter CVCC ICC + Daux Laux Vtimer 5 FB R1 2.5k C1 100n IFB < 40 mA ? = Low Else = High Auto-Recovery - B Version VCC Itimer 10m Timer Ctimer 100nF VCC Pon Reset FB C1 100n R1 2.5k IFB DFB IFB IFB + + VCC + - 20ms Filter 6V CVCC ICC + Daux Laux Vtimer 5 - + SCR Delatches When ISCR < ICClatch (Few mA) IFB < 40 mA ? = Low Else = High Latched - A Version Figure 17. The Internal Fault Management Differs Depending on the Considered Version Knowing both the ending voltage and the charge current, we can easily calculate the timer capacitor value for a given delay. Suppose we need 40 ms. In that case, the capacitor is simply: 11.7 m 40 m I T Ctimer + timer + + 94 nF 5 Vtimer (eq. 15) the target). Yes, you can use this reference voltage to supply a NTC and form a cheap OTP protection. VCC Select a 100 nF value. Latch Input 5V C2 100n FB Latch OVP D2 The NCP1351 features a patented circuitry which prevents the FB input to be of low impedance before the Vcc reaches the VCCON level. As such, the circuit can work in a primary regulation scheme. Capitalizing on this typical option, Figure 18 shows how to insert a zener diode in series with the optocoupler emitter pin. In that way, the current biases the zener diode and offers a nice reference voltage, appearing at the loop closure (e.g. when the output reaches R1 2.5k C1 100nF Rpulldown C3 100nF Figure 18. The Latch Input Offers Everything Needed to Implement an OTP Circuit. Another Zener Can Help combining an OVP Circuit if Necessary http://onsemi.com 14 NCP1351 VCC VCC Aux + OUT + CVCC 20mF Laux R4 2.2k CVCC 22mF Sec U1A U1B Latch Rpulldown ROVP C3 100nF D2 1N4937 Latch C5 1n C1 100nF D4 C4 100n Figure 19. You can either directly observe the VCC level or add a small RC filter to reduce the leakage inductance contribution. The best is to directly sense the output voltage and reacts if it runs away, as offered on the right side. Design Example, a 19 V / 3 Solving for N gives: N+ kC Vout ) Vf 1.6 Ns + + Np Vclamp + 0.234 (19 ) 0.8) 135 (eq. 19) A Universal Mains Power Supply Designing a Switch-Mode Power Supply using the NCP1351 does not differ from a fixed frequency design. What changes, however, is the regulation method via frequency variations. In other words, all the calculations must be carried at the lowest line input where the frequency will hit the maximum value set by the Ct capacitor. Let us follow the steps: Vin min = 100 Vdc (bulk valley in low-line conditions) Vin max = 375 Vdc Vout = 19 V Iout = 3 A Operating mode is CCM = 0.8 Fsw = 65 kHz 1. Turn Ratio. This is the first parameter to consider. The MOSFET BVdss actually dictates the amount of reflected voltage you need. If we consider a 600 V MOSFET and a 15% derating factor, we must limit the maximum drain voltage to: Vds_max + 600 0.85 + 510 V (eq. 16) Let us round it to 0.25 or 1/N = 4 Ipeak I1 Ivalley DIL Iavg Knowing a maximum bulk voltage of 375 V, the clamp voltage must be set to: Vclamp + 510 * 375 + 135 V (eq. 17) t DTSW TSW Based on the above level, we decide to adopt a headroom between the reflected voltage and the clamp level of 50 V. If this headroom is too small, a high dissipation will occur on the RDC clamp network and efficiency will suffer. A leakage inductance of around 1% of the magnetizing value should give good results with this choice (kc = 1.6). The turn ratio between primary and secondary is simply: Vout ) Vf N + Vclamp kc (eq. 18) Figure 20. Primary Inductance Current Evolution in CCM 2. Calculate the maximum operating duty-cycle for this flyback converter operated in CCM: d max + Vout N Vout N ) Vin_min + 19 19 4 + 0.43 4 ) 100 (eq. 20) http://onsemi.com 15 NCP1351 In this equation, the CCM duty-cycle does not exceed 50%. The design should thus be free of subharmonic oscillations in steady-state conditions. If necessary, negative ramp compensation is however feasible by the auxiliary winding. 3. To obtain the primary inductance, we can use the following equation which expresses the inductance in relationship to a coefficient k. This coefficient actually dictates the depth of the CCM operation. If it goes to 2, then we are in DCM. L+ (Vin_mind max)2 FSWKPin + 1.65 + 1.1 A (eq. 21) On Figure 20, I1 can also be calculated: II + Ipeak * 1.34 DIL + 2.33 * + 1.65 A 2 2 (eq. 26) The valley current is also found to be: Ivalley + Ipeak * DIL + 2.33 * u1.34 + 1.0 A (eq. 27) 4. Based on the above numbers, we can now evaluate the RMS current circulating in the MOSFET and the sense resistor: Id_rms + II d 1) 1 DIL 2 3 2I1 1) 1.34 2 1 3 2 1.65 (eq. 28) where K = DIL/II and defines the amount of ripple we want in CCM (see Figure 20). * Small K: deep CCM, implying a large primary inductance, a low bandwidth and a large leakage inductance. * Large K: approaching BCM where the RMS losses are the worse, but smaller inductance, leading to a better leakage inductance. From Equation 16, a K factor of 0.8 (40% ripple) ensures a good operation over universal mains. It leads to an inductance of: L+ (100 65 k 43)2 0.8 72 + 493 mH 3 100 (eq. 22) 0.65 5. The current peaks to 2.33 A. Selecting a 1 V drop across the sense resistor, we can compute its value: Rsense + 1 Ipeak + 1 + 0.4 W 2.5 (eq. 29) To generate 1 V, the offset resistor will be 3.7 kW, as already explained. Using Equation 28, the power dissipated in the sense element reaches: Psense + Rsense Id_rms2 + 0.4 1.12 + 484 mW (eq. 30) Vin_mind max 19 DIL + + 0.8 LFSW + 1.34 A peak-to-peak (eq. 23) The peak current can be evaluated to be: Iin_avg + Pout 100 + 493 m hVin_min ) 0.43 65 k + 712 mA (eq. 24) 6. To switch at 65 kHz, the Ct capacitor connected to pin 2 will be selected to 180 pF. 7. As the load changes, the operating frequency will automatically adjust to satisfy either equation 5 (high power, CCM) or equation 6 in lighter load conditions (DCM). Figure 21 portrays a possible application schematic implementing what we discussed in the above lines. Ipeak + Iavg d 0.712 1.34 DIL + ) + 2.33 A 0.43 2 2 (eq. 25) http://onsemi.com 16 NCP1351 HV-Bulk R3 47k R13 47k R4 22 C2 10n 400V LP = 500mH NP:NS = 1:0.25 NP:Naux = 0.18 D5 MBR20200 C5a 1.2mF 25V C13 2.2nF Type = Y1 T1 + R7 1M R2 1M D2 MUR 160 8 7 6 5 NCP1351B OVP Option C5b 1.2mF 25V + L2 2.2m C7 220mF 25V + VOUT 19V/3A D3 1N4937 U1B 1 C12 + 100mF 400V R15 3.7k C15 22p 2 3 4 U2 D6 1N4148 R8 1k GND R12 4k 25V 6A/600V M1 R10 62k R16 10 R18 47k + C17 100m U1A R14 2.2k C6 100n C4 100n R5 2.5k C9 100n C8 270pF R1 2.2k C10 0.1m + IC2 TL431 R9 10k R6 0.4 C3 C1 4.7m 100nF 25V GND Figure 21. The 19 V Adapter Featuring the Elements Calculated Above On this circuit, the VCC capacitor is split in two parts, a low value capacitor (4.7 mF) and a bigger one (100 mF). The 4.7 mF capacitor ensures a low startup time, whereas the second capacitor keeps the VCC alive in standby mode (where the switching frequency can be low). Due to D6, it does not hamper startup time. Application Results Another benefit of the variable frequency lies in the low ripple operation at no-load. This is what confirms Figure 23. Finally, the power supply was tested for its transient response, from 100 mA to 3 A, high and low line, with a slew-rate of 1 A/ms (Figure 25). Results appear in Figures 25 and 26 and confirm the stability of the board. We assembled a board with component values close to what is described on Figure 21. Here are the obtained results: Pin @ no-load = 152 mW, Vin = 230 Vac Pin @ no-load = 164 mW, Vin = 100 Vac The efficiency stays flat to above 80%, and keeps good even at low output levels. It clearly shows the benefit of the variable frequency implemented in the NCP1351. 88 86 84 EFFICIENCY (%) 82 80 78 76 74 72 0 0.5 1 1.5 2 Iout (A) 2.5 3 3.5 Vin = 100 Vac Vin = 230 Vac Figure 22. Efficiency Measured at Various Operating Points http://onsemi.com 17 NCP1351 Vds 200 V/div Vds 200 V/div Vout 1.0 mV/div Vout 1.0 mV/div Figure 23. No-Load Output Ripple (Vin = 230 Vac) Figure 24. Same Conditions, Pout = 5 W Vout 50 mV/div Vout 50 mV/div Figure 25. Transient Step, Low Line Figure 26. Transient Step, High Line http://onsemi.com 18 NCP1351 PACKAGE DIMENSIONS SOIC-8 D SUFFIX CASE 751-07 ISSUE AH -X- A 8 5 NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: MILLIMETER. 3. DIMENSION A AND B DO NOT INCLUDE MOLD PROTRUSION. 4. MAXIMUM MOLD PROTRUSION 0.15 (0.006) PER SIDE. 5. DIMENSION D DOES NOT INCLUDE DAMBAR PROTRUSION. ALLOWABLE DAMBAR PROTRUSION SHALL BE 0.127 (0.005) TOTAL IN EXCESS OF THE D DIMENSION AT MAXIMUM MATERIAL CONDITION. 6. 751-01 THRU 751-06 ARE OBSOLETE. NEW STANDARD IS 751-07. MILLIMETERS MIN MAX 4.80 5.00 3.80 4.00 1.35 1.75 0.33 0.51 1.27 BSC 0.10 0.25 0.19 0.25 0.40 1.27 0_ 8_ 0.25 0.50 5.80 6.20 INCHES MIN MAX 0.189 0.197 0.150 0.157 0.053 0.069 0.013 0.020 0.050 BSC 0.004 0.010 0.007 0.010 0.016 0.050 0_ 8_ 0.010 0.020 0.228 0.244 B 1 4 S 0.25 (0.010) M Y M -Y- G C -Z- H D 0.25 (0.010) M SEATING PLANE K N X 45 _ 0.10 (0.004) M J ZY S X S DIM A B C D G H J K M N S SOLDERING FOOTPRINT* 1.52 0.060 7.0 0.275 4.0 0.155 0.6 0.024 1.270 0.050 SCALE 6:1 mm inches *For additional information on our Pb-Free strategy and soldering details, please download the ON Semiconductor Soldering and Mounting Techniques Reference Manual, SOLDERRM/D. ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. "Typical" parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including "Typicals" must be validated for each customer application by customer's technical experts. SCILLC does not convey any license under its patent rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner. PUBLICATION ORDERING INFORMATION LITERATURE FULFILLMENT: Literature Distribution Center for ON Semiconductor P.O. Box 5163, Denver, Colorado 80217 USA Phone: 303-675-2175 or 800-344-3860 Toll Free USA/Canada Fax: 303-675-2176 or 800-344-3867 Toll Free USA/Canada Email: orderlit@onsemi.com N. American Technical Support: 800-282-9855 Toll Free USA/Canada Europe, Middle East and Africa Technical Support: Phone: 421 33 790 2910 Japan Customer Focus Center Phone: 81-3-5773-3850 ON Semiconductor Website: www.onsemi.com Order Literature: http://www.onsemi.com/orderlit For additional information, please contact your local Sales Representative http://onsemi.com 19 NCP1351/D |
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